前沿方案延缓LED老化

改进光耦电路，可以提升零电压测量的精度。

　　有不同潜路工作的电路中，用光耦建立电流隔离看上去似乎很简单。光耦从隔离电路中获取能量，由于LED老化，开关相对慢且不稳定。若不用光耦，可使用如Analog Devices公司的ADUM12xx或Texas Instruments 公司的ISO72x替代。本设计方案阐述了一个简单改进光耦电路的方法。

　　图1显示了两个通用的0V同步交流设计。通过光耦负载电阻的减少，开关变得更慢更不确定，但减少了光耦的LED电流，尝试减少隔离电路中的能量消耗。为实现更快更迅速的开关，将不得不牺牲能量效率；然而，由于能量效率和交流电压大小的反向关系，这个牺牲的好处是有限的。

　　光耦的LED在近似全交流循环过程中超乎寻常的几乎连续发光，导致功耗效率低，且使光耦老化得相对较快：一个显著的缺点是过原点误差过大且几乎不可控；电路的灵敏度范围依靠光耦的参数。图1的设计不是一个理想方案。就效率而言，依靠光耦的电流转换率和交流幅值，它们能输出5到100 mA。

　　图2的设计克服了能耗过大、不确定开关和LED老化的问题。它非常适用于宽交流范围的应用。与图1的电路相比，图2的LED只在过原点附近发光，且由前置充电电容接收能量，所以通过10到100的因数减少平均电流消耗。设计也提供更快、更确定和更敏锐的开关。更甚者，希望延缓LED老化。图1中电阻R1和R2消耗的热功率不小于1.5W，所以在同一电路板区域用0.1W设备替换外部器件（图2）。

　　电路的主要部分由幅值检波器D1、电容C1和Schmitt触发器Q1/Q2组成，控制流过光耦的LED电流。D2和D3稳定Q2的基电压，从而其集电极电流驱动光耦。电容C1通过R1、R2和D1充电。

　　几乎所有交流周期中，除了过原点附近，Q1为开，Q2为关。然后，接近过原点时，
Schmitt触发器Q1和Q2的状态改变，Q2以恒定的电流卸放电容C1，因为由Q2、D2、D3、R5和R6组成的电路按I=(2×VD–VBE2)/R6稳定电流，在这里VD为D2或D3上的电压降，VBE2为Q2的基射极电压。

　　一些应用不需要Schmitt触发器固有的磁滞性；图3显示了这样的一个设计。它也显示了怎样处理不需要的D1最小反转电流。然而，电路更适用于纯同步和非晶闸管控制。由于LED电流的稳定性，这些设计使输入交流电压的范围扩大，其有利于多标准交流供电设计；有机会在LED没有过载危险的情况下设置LED电流；减少光耦不稳定的影响。这样设计的另一个优势为其固有更安全的特性。在其终端短路的情况下，光耦在隔离与非隔离侧之间传递的电流比图1电路中少10到100倍。光耦也有优势。由于低占空比，可以不损失功率而任意减少光耦负载电阻R8的值。这个减少将使过原点误差降低。

　　英文原文：

　　Improved optocoupler circuits reduce current draw, resist LED aging

　　With improved optocoupler circuits, you can enhance the accuracy of zero-voltage sensing.

　　Peter Demchenko, Vilnius, Lithuania; Edited by Charles H Small and Fran Granville -- EDN, 12/14/2007

　　It seems deceptively simple to establish galvanic isolation with the help of optocouplers between circuits that operate at different ground potentials. Optocouplers draw power from the isolated circuit, and switching can be relatively slow and uncertain because of LED aging. Substitutes without optocouplers, such as the ADUM12xx from Analog Devices or ISO72x from Texas Instruments, are available. This Design Idea describes a method of improving the simple optocoupler.

　Figure 1 shows two popular designs of 0V synchronization with ac. Anattempt to reduce power draw from the isolated circuit by decreasing the optocoupler’s LED current with a corresponding increase of the optocoupler’s load resistor yields slower and more uncertain switching. To achieve faster and sharper switching, you would have to sacrifice power efficiency; however, the benefit of this sacrifice is limited because of the inverse relationship between power efficiency and the ac-voltage magnitude.

　　An optocoupler’s LED emits almost continuously during nearly all ac cycles exceeding the nominal, leading to low power efficiency and relatively fast aging of the optocoupler. One more drawback is excessively large and nearly uncontrollable zero-crossing error; the circuit’s sensitivity threshold depends on the parameters of the optocoupler. The designs in Figure 1 do not provide an ideal approach. With respect to efficiency, they can draw 5 to 100 mA, depending on the optocoupler’s current-transfer ratio and the ac amplitude.

　　The design in Figure 2 overcomes the problems of excessive power consumption, uncertain switching, and LED aging. It lends itself well to wide-ac-range applications. Compared with the circuit in Figure 1, Figure 2’s LED emits only in close vicinity of the zero-crossing point and receives its power from the previously charged capacitor, so you can reduce the average current draw by a factor of 10 to 100. The design also provides faster, more deterministic, and sharper switching. What’s more, you can expect slower LED aging. Resistors R1 and R2 in Figure 1 dissipate no less than 1.5W of power as waste heat, so changing them to 0.1W devices allows placement of additional components on the same board area (Figure 2).

　　The circuit’s main components comprise amplitude detector D1, capacitor C1, and Schmitt trigger Q1/Q2 to control a current through the optocoupler’s LED. D2 and D3 stabilize the base voltage of Q2 and, hence, its collector current, which activates the optocoupler. Capacitor C1 charges up through R1, R2, and D1.

　　During nearly all of the ac-cycle time, except in the vicinity of the zero-crossing point, Q1 is on, and Q2 is off. Then, approaching the zero-crossing point, the state of Schmitt trigger Q1 and Q2 changes, and Q2 discharges capacitor C1 with the constant current, because the circuit comprising Q2, D2, D3, R5, and R6 stabilizes current as I=(2×VD–VBE2)/R6, where VD is the voltage drop on D2 or D3 and VBE2 is the base-emitter voltage of Q2.

　　Some applications require none of
the hysteresis that is inherent to a Schmitt trigger; Figure 3 shows such a design. It also shows how to manage without a requirement for minimal reverse current in D1. This circuit, however, better suits pure synchronization and not thyristor control. Because of the stability of LED current, these designs provide an expanded input-ac-voltage range, which may be useful for a multistandard ac-powered gadget; an opportunity to set the LED current without the risk of overloading the LED; and a reduced influence of the optocoupler’s instability. One more advantage of these designs is their inherently safer nature. In the case of a short circuit in their terminals, optocouplers deliver 10 to 100 times less current between the isolated and the nonisolated sides than the circuit in Figure 1. The optocoupler also offers advantages. Thanks to the low duty cycle, you can freely reduce the value of the optocoupler’s load resistor, R8, without sacrificing power efficiency. This reduction results in low zero-crossing error.

　　英文原文地址：http://www.edn.com/article/CA6512152.html